In the 1980's, the United States Federal Communications Commission (FCC) adopted new regulations covering the audio portion of television signals which permitted television programs to be broadcast and received with bichannel audio, e.g., stereophonic sound. In those regulations, the FCC recognized and gave special protection to a method of broadcasting additional audio channels endorsed by the Electronic Industries Association and the National Association of Broadcasters and called the Broadcast Television Systems Committee (BTSC) system. This well known standard is sometimes referred to as Multichannel Television Sound (MTS) and is described in the FCC document entitled, MULTICHANNEL TELEVISION SOUND TRANSMISSION AND AUDIO PROCESSING REQUIREMENTS FOR THE BTSC SYSTEM (OET Bulletin No. 60, Revision A, February 1986), as well as in the document published by the Electronic Industries Association entitled, MULTICHANNEL TELEVISION SOUND BTSC SYSTEM RECOMMENDED PRACTICES (EIA Television Systems Bulletin No. 5, July 1985). Television signals generated according to the BTSC standard are referred to hereinafter as “BTSC signals”.
The original monophonic television signals carried only a single channel of audio. Due to the configuration of the monophonic television signal and the need to maintain compatibility with existing television sets, the stereophonic information was necessarily located in a higher frequency region of the BTSC signal making the stereophonic channel much noisier than the monophonic audio channel. This resulted in an inherently higher noise floor for the stereo signal than for the monophonic signal. The BTSC standard overcame this problem by defining an encoding system that provided additional signal processing for the stereophonic audio signal. Prior to broadcast of a BTSC signal by a television station, the audio portion of a television program is encoded in the manner prescribed by the BTSC standard, and upon reception of a BTSC signal a receiver (e.g., a television set) then decodes the audio portion in a complementary manner. This complementary encoding and decoding insures that the signal-to-noise ratio of the entire stereo audio signal is maintained at acceptable levels.
FIG. 1 is a block diagram of a prior art BTSC encoding system, or more simply, a BTSC encoder 100, as defined by the BTSC standard. Encoder 100 receives left and right channel audio input signals (indicated in FIG. 1 as “L” and “R”, respectively) and generates therefrom a conditioned sum signal and an encoded difference signal. It should be appreciated that while the system of the prior art and that of the present invention is described as useful for encoding the left and right audio signals of a stereophonic signal that is subsequently transmitted as a television signal, the BTSC system also provides means to encode a separate audio signal, e.g., audio information in a different language, which is separated and selected by the end receiver. Further, noise reduction components of the BTSC encoding system can be used for other purposes besides television broadcast, such as for improving audio recordings.
System 100 includes an input section 110, a sum channel processing section 120, and a difference channel processing section 130. Input section 110 receives the left and right channel audio input signals and generates therefrom a sum signal (indicated in FIG. 1 as “L+R”) and a difference signal (indicated in FIG. 1 as “L−R”). It is well known that for stereophonic signals, the sum signal L+R may be used by itself to provide monophonic audio reproduction and it is this signal that is decoded by existing monophonic audio television sets to reproduce sound. In stereophonic sets, the sum and difference signals can be added to and subtracted from one another to recover the original two stereophonic signals (L) and (R). Input section 110 includes two signal adders 112, 114. Adder 112 sums the left and right channel audio input signals to generate the sum signal, and adder 114 subtracts the right channel audio input signal from the left channel audio input signal to generate the difference signal. As described above, the sum signal L+R is transmitted through a transmission media with the same signal to noise ratio as achieved with the prior monophonic signals. However, the difference signal L−R is transmitted though a very noisy channel, particularly at the higher frequency portion of the relevant spectrum so that the decoded difference signal has a poorer signal-to-noise ratio because of the noisy medium and reduced dynamic range of the medium. The dynamic range is defined as the range of signals between the level of the noise floor and the maximum level where signal saturation occurs. In the difference signal channel the dynamic range decreases at higher frequencies. Accordingly, the difference signal is subjected to additional processing than that of the sum signal so that the dynamic range can be substantially preserved.
More particularly, the sum channel processing section 120 receives the sum signal and generates therefrom the conditioned sum signal. Section 120 includes a 75 μs preemphasis filter 122 and a bandlimiter 124. The sum signal is applied to the input of filter 122 which generates therefrom an output signal that is applied to the input of bandlimiter 124. The output signal generated by the latter is then the conditioned sum signal.
The difference channel processing section 130 receives the difference signal and generates therefrom the encoded difference signal. Section 130 includes a fixed preemphasis filter 132 (shown implemented as a cascade of two filters 132a and 132b), a variable gain amplifier 134 preferably in the form of a voltage-controlled amplifier, a variable preemphasis/deemphasis filter (referred to hereinafter as a “variable emphasis filter”) 136, an overmodulation protector and bandlimiter 138, a fixed gain amplifier 140, a bandpass filter 142, an RMS level detector 144, a fixed gain amplifier 146, a bandpass filter 148, an RMS level detector 150, and a reciprocal generator 152.
The difference signal is applied to the input of fixed preemphasis filter 132 which generates therefrom an output signal that is applied via line 132d to an input terminal of amplifier 134. An output signal generated by reciprocal generator 152 is applied via line 152a to a gain control terminal of amplifier 134. Amplifier 134 generates an output signal by amplifying the signal on line 132d using a gain that is proportional to the value of the signal on line 152a. The output signal generated by amplifier 134 is applied via line 134a to an input terminal of variable emphasis filter 136, and an output signal generated by RMS detector 144 is applied via line 144a to a control terminal of filter 136. Variable emphasis filter 136 generates an output signal by preemphasizing or deemphasizing the high frequency portions of the signal on line 134a under the control of the signal on line 144a. The output signal generated by filter 136 is applied to the input of overmodulation protector and bandlimiter 138 which generates therefrom the encoded difference signal.
The encoded difference signal is applied via feedback path 138a to the inputs of fixed gain amplifiers 140, 146, which amplify the encoded difference signal by Gain A and Gain B, respectively. The amplified signal generated by amplifier 140 is applied to an input of bandpass filter 142 which generates therefrom an output signal that is applied to the input of RMS level detector 144. The latter generates an output signal as a function of the RMS value of the input signal level received from filter 142. The amplified signal generated by amplifier 146 is applied to the input of bandpass filter 148 which generates therefrom an output signal that is applied to the input of RMS level detector 150. The latter generates an output signal as a function of the RMS value of the input signal level received from filter 148. The output signal of detector 150 is applied via line 150a to reciprocal generator 152, which generates a signal on line 152a that is representative of the reciprocal of the value of the signal on line 150a. As stated above, the output signals generated by RMS level detector 144 and reciprocal generator 152 are applied to filter 136 and amplifier 134, respectively.
As shown in FIG. 1, the difference channel processing section 130 is considerably more complex than the sum channel processing section 120. The additional processing provided by the difference channel processing section 130, in combination with complementary processing provided by a decoder (not shown) receiving a BTSC signal, maintains the signal-to-noise ratio of the difference channel at acceptable levels even in the presence of the higher noise floor associated with the transmission and reception of the difference channel. Difference channel processing section 130 essentially generates the encoded difference signal by dm compressing, or reducing the dynamic range of the difference signal so that the encoded signal may be transmitted through the limited dynamic range transmission path associated with a BTSC signal, and so that a decoder receiving the encoded signal may recover all the dynamic range in the original difference signal by expanding the compressed difference signal in a complementary fashion. The difference channel processing section 130 is a particular form of the adaptive signal weighing system described in U.S. Pat. No. 4,539,526, which is known to be advantageous for transmitting a signal having a relatively large dynamic range through a transmission path having a relatively narrow, frequency dependent, dynamic range.
Briefly, the difference channel processing section may be thought of as including a wide band compression unit 180 and a spectral compression unit 190. The wide band compression unit 180 includes variable gain amplifier 134 preferably in the form of a voltage controlled amplifier, and the components of the feedback path for generating the control signal to amplifier 134 and comprising amplifier 146, band pass filter 148, RMS level detector 150, and reciprocal generator 152. Band pass filter 148 has a relatively wide pass band, weighted towards lower audio frequencies, so in operation the output signal generated by filter 148 and applied to RMS level detector 150 is substantially representative of the encoded difference signal. RMS level detector 150 therefore generates an output signal on line 150a representative of a weighted average of the energy level of the encoded difference signal, and reciprocal generator 152 generates a signal on line 152a representative of the reciprocal of this weighted average. The signal on line 152a controls the gain of amplifier 134, and since this gain is inversely proportional to a weighted average (i.e., weighted towards lower audio frequencies) of the energy level of the encoded difference signal, wide band compression unit 180 “compresses”, or reduces the dynamic range, of the signal on line 132a by amplifying signals having relatively low amplitudes and attenuating signals having relatively large amplitudes.
The spectral compression unit 190 includes variable emphasis filter 136 and the components of the feedback path generating a control signal to the filter 136 and comprising amplifier 140, band pass filter 142 and RMS level detector 144. Unlike filter 148, band pass filter 142 has a relatively narrow pass band that is weighted towards higher audio frequencies. As is well known, the transmission medium associated with the difference portion of the BTSC transmission system has a frequency dependent dynamic range and the pass band of filter 142 is chosen to correspond to the spectral portion of that transmission path having the narrowest dynamic range (i.e., the higher frequency portion). In operation the output signal generated by filter 142 and applied to RMS level detector 144 contains primarily the high frequency portions of the encoded difference signal. RMS level detector 144 therefore generates an output signal on line 144a representative of the energy level in the high frequency portions of the encoded difference signal. This signal then controls the preemphasis/deemphasis applied by variable emphasis filter 136 so in effect the spectral compression unit 190 dynamically compresses high frequency portions of the signal on line 134a by an amount determined by the energy level in the high frequency portions of the encoded difference signal as determined by the filter 142. The use of the spectral compression unit 190 thus provides additional signal compression towards the higher frequency portions of the difference signal, which combines with the wideband compression provided by the variable gain amplifier 134 to effectively cause more overall compression to take place at high frequencies relative to the compression at lower frequencies. This is done because the difference signal tends to be noisier in the higher frequency part of the spectrum. When the encoded difference signal is decoded with a wideband expander and a spectral expander in a decoder (not shown), respectively in a complementary manner to the wide band compression unit 180 and spectral compression unit 190 of the encoder, the signal-to-noise ratio of the L−R signal applied to the difference channel processing section 130 will be substantially preserved.
The BTSC standard rigorously defines the desired operation of the 75 μs preemphasis filter 122, the fixed preemphasis filter 132, the variable emphasis filter 136, and the bandpass filters 142, 148, in terms of idealized analog filters. Specifically, the BTSC standard provides a transfer function for each of these components and the transfer functions are described in terms of mathematical representations of idealized analog filters. The BTSC standard also defines the gain settings, Gain A and Gain B, of amplifiers 140 and 146, respectively, and also defines the operation of amplifier 134, RMS level detectors 144, 150, and reciprocal generator 152. The BTSC standard also provides suggested guidelines for the operation of overmodulation protector and bandlimiter 138 and bandlimiter 124. Specifically, bandlimiter 124 and the bandlimiter portion of overmodulation protector and bandlimiter 138 are described as low pass filters with cutoff frequencies of 15 kHz, and the overmodulation protection portion of overmodulation protector and bandlimiter 138 is described as a threshold device that limits the amplitude of the encoded difference signal to 100% of full modulation where full modulation is the maximum permissible deviation level for modulating the audio subcarrier in a television signal.
Since encoder 100 is defined in terms of mathematical descriptions of idealized filters it may be thought of as an idealized or theoretical encoder, and those skilled in the art will appreciate that it is virtually impossible to construct a physical realization of a BTSC encoder that exactly matches the performance of theoretical encoder 100. Therefore, it is expected that the performance of all BTSC encoders will deviate somewhat from the theoretical ideal, and the BTSC standard defines maximum limits on the acceptable amounts of deviation. For example, the BTSC standard states that a BTSC encoder must provide at least 30 db of separation from 100 Hz to 8,000 Hz where separation is a measure of how much a signal applied to only one of the left or right channel's inputs appears erroneously in the other of the left or right channel's outputs.
The BTSC standard also defines a composite stereophonic baseband signal (referred to hereinafter as the “composite signal”) that is used to generate the audio portion of a BTSC signal. The composite signal is generated using the conditioned sum signal, the encoded difference signal, and a tone signal, commonly referred to as the “pilot tone” or simply as the “pilot”, which is a sine wave at a frequency fH where fH is equal to 15,734 Hz. The presence of the pilot in a received television signal indicates to the receiver that the television signal is a BTSC signal rather than a monophonic or other non BTSC signal. The composite signal is generated by multiplying the encoded difference signal by a waveform that oscillates at twice the pilot frequency according to the cosine function cos(4πfHt), where t is time, to generate an amplitude modulated, double-sideband, suppressed carrier signal and by then adding to this signal the conditioned sum signal and the pilot tone.
FIG. 2 is a graph of the spectrum of the composite signal. In FIG. 2 the spectral band of interest containing the content of the conditioned sum signal (or the “sum channel signal”) is indicated as “L+R”, the two spectral sidebands containing the content of the frequency shifted encoded difference signal (or the “difference channel signal”) are each indicated as “L−R”, and the pilot tone is indicated by the arrow at frequency fH. As shown in FIG. 2, in the composite signal the encoded difference signal is used at 100% of full modulation, the conditioned sum signal is used at 50% of full modulation, and the pilot tone is used at 10% of full modulation.
Stereophonic television has been widely successful, and existing encoders have performed admirably, however, virtually every BTSC encoder now in use has been built using analog circuitry technology. These analog BTSC encoders, and particularly the analog difference channel processing sections, due to their increased complexity have been relatively difficult and expensive to construct. Due to the variability of analog components, complex component selection and extensive calibration have been required to produce acceptable analog difference channel processing sections. Further, the tendency of analog components to drift, over time, away from their calibrated operating points has also made it difficult to produce an analog difference channel processing section that consistently and repeatably performs within a given tolerance. A digital difference channel processing section, if one could be built, would not suffer from these problems of component selection, calibration, and performance drift, and could potentially provide increased performance.
Further, the analog nature of existing BTSC encoders has made them inconvenient to use with newly developed, increasingly popular, digital equipment. For example, television programs can now be stored using digital storage media such as a hard disk or digital tape, rather than the traditional analog storage media, and in the future increasing use will be made of digital storage media. Generating a BTSC signal from a digitally stored program now requires converting the digital audio signals to analog signals and then applying the analog signals to an analog BTSC encoder. A digital BTSC encoder, if one could be built, could accept the digital audio signals directly and could therefore be more easily integrated with other digital equipment.
While a digital BTSC encoder would potentially offer several advantages, there is no simple way to construct an encoder using digital technology that is functionally equivalent to the idealized encoder 100 defined by the BTSC standard. One problem is that the BTSC standard defines all the critical components of idealized encoder 100 in terms of analog filter transfer functions. As is well known, while it is generally possible to design a digital filter so that either the magnitude or the phase response of the digital filter matches that of an analog filter, it is extremely difficult to match both the amplitude and phase responses without requiring large amounts of processing capacity for processing data sampled at very high sampling rates or without significantly increasing the complexity of the digital filter. Without increasing either the sampling frequency or the filter order, the amplitude response of a digital filter can normally only be made to more closely match that of an analog filter at the expense of increasing the disparity between the phase responses of the two filters, and vice versa. However, since small errors in either amplitude or phase decrease the amount of separation provided by BTSC encoders, it would be essential for a digital BTSC encoder to closely match both the amplitude and phase responses of an idealized encoder of the type shown at 100 in FIG. 1.
For a digital BTSC encoder to provide acceptable performance, it is critical to preserve the characteristics of the analog filters of an idealized encoder 100. Various techniques exist for designing a digital filter to match the performance of an analog filter; however, in general, none of these techniques produce a digital filter (of the same order as the analog filter) having amplitude and phase responses that exactly match the corresponding responses of the analog filter. Ideal encoder 100 is defined in terms of analog transfer functions specified in the frequency domain, or the s-plane, and to design a digital BTSC encoder, these transfer functions must be transformed to the z-plane. Such a transformation may be performed as a “many-to-one” mapping from the s-plane to the z-plane which attempts to preserve time domain characteristics. However, in such a transformation the frequency domain responses are subject to aliasing and may be altered significantly. Alternatively, the transformation may be performed as a “one-to-one” mapping from the s-plane to the z-plane that compresses the entire s-plane into the unit circle of the z-plane. However, such a compression suffers from the familiar “frequency warping” between the analog and digital frequencies. Prewarping can be employed to compensate for this frequency warping effect, however, prewarping does not completely eliminate the deviations from the desired frequency response. These problems would have to be overcome to produce a digital BTSC encoder that performs well and is not unduly complex or expensive.
There is therefore a need for overcoming the difficulties and developing a digital BTSC encoder.